Adaptive equalization using correlation of data patterns with errors

ABSTRACT

An integrated receiver supports adaptive receive equalization. An incoming bit stream is sampled using edge and data clock signals derived from a reference clock signal. A phase detector determines whether the edge and data clock signals are in phase with the incoming data, while some clock recovery circuitry adjusts the edge and data clock signals as required to match their phases to the incoming data. The receiver employs the edge and data samples used to recover the edge and data clock signals to note the locations of zero crossings for one or more selected data patterns. The pattern or patterns may be selected from among those apt to produce the greatest timing error. Equalization settings may then be adjusted to align the zero crossings of the selected data patterns with the recovered edge clock signal.

FIELD OF THE INVENTION

The present invention relates generally to the field of communications,and more particularly to high-speed electronic signaling within andbetween integrated circuit devices.

BACKGROUND

Communication channels typically exhibit low pass filter effects thatdisproportionately attenuate high-frequency signal components. Theseeffects can vary from one channel to the next, and can vary over time ina given channel. Adaptive receive equalization schemes are thereforeused in high-speed communication links to compensate for all or part ofthe distortion imposed by the channel.

The amount of channel-induced distortion appearing on any particular bitin a serial data signal is pattern dependent. This pattern dependencyowes to the fact that different data patterns have different spectralcontent, and are thus affected differently by the channel transferfunction. As a first-order approximation for a typical channel, thehigher the frequency, the greater the attenuation.

Equalization refers generally to processes for emphasizing orattenuating a selected frequency or frequencies of a signal, often tocompensate for frequency-specific attenuation of the signal.Equalization schemes can be “adaptive,” in which case the equalizationparameters may be dynamically adjusted to account for variables thataffect the communication channel, including process variations andfluctuations in temperature, supply voltage, and the noise environment.Many of these adaptive equalization schemes require sensitive analogcircuitry and/or additional samplers that significantly increase systemcomplexity, implementation difficulty, and power requirements. There istherefore a need for efficient adaptive receiver equalization systemsand methods that are more easily implemented and verified, with reducedpower penalty.

BRIEF DESCRIPTION OF THE DRAWINGS

The subject matter disclosed is illustrated by way of example, and notby way of limitation, in the figures of the accompanying drawings and inwhich like reference numerals refer to similar elements and in which:

FIG. 1 depicts an integrated receiver 100 that supports adaptive receiveequalization in accordance with one embodiment.

FIGS. 2 and 3 depict unequalized input data signals for two datapatterns, (10101) and (11101), respectively, to illustrate how differentdata patterns suffer different levels of distortion.

FIG. 4 is a flowchart 400 describing an adaptive equalization methodcarried out by receiver 100 of FIG. 1 .

FIG. 5 is a waveform diagram illustrating an unequalized waveform 300and a corresponding equalized version of waveform 300 as a waveform 505.

FIG. 6 depicts waveforms 200 and 500, of FIGS. 2 and 5 respectively,overlaid to show coincident zero crossings for the two waveforms.

FIG. 7 depicts a double-data-rate (DDR) communication system 700 inaccordance with another embodiment.

FIG. 8 depicts pattern mask 755 and equalization logic 760, both of FIG.7 , in accordance with one embodiment.

FIG. 9 is a waveform diagram 900 illustrating the operation of patternmask 755 and the early and late registers 815 and 820 upon receipt of aten-bit data word.

FIG. 10 is a flowchart 1002 depicting a method of operation for thecircuitry of FIG. 8 in accordance with one embodiment.

FIGS. 11 and 12 are waveform diagrams depicting a pair of data waveforms1105 and 1115, representing respective received data patterns (11101)and (00001).

FIG. 13 depicts a DDR receiver 1300 in accordance with anotherembodiment. Receiver 1300 includes an equalizer 1325 that equalizes adifferential input data signal Vin_p/Vin_n to produce an equalizedsignal VEQ.

FIG. 14 depicts pattern mask 755 of FIG. 7 and some equalization logic1400 that together adjust equalization signal EQ[3:0] responsive to upto four mask patterns.

FIG. 15 is a flowchart 1500 depicting the operation of equalizationlogic 1400 of FIG. 14 in accordance with one embodiment. The logic offlowchart 1500 is described here in connection with FIG. 14 .

DETAILED DESCRIPTION

FIG. 1 depicts an integrated receiver 100 that supports adaptiveequalization in accordance with one embodiment. As is typical in serialreceiver architectures, an incoming bit stream is sampled using edge anddata clock signals derived from a reference clock signal. A phasedetector determines whether the edge and data clock signals are in phasewith the incoming data, while some clock recovery circuitry adjusts theedge and data clock signals as required to match their phases to theincoming data. In accordance with the depicted embodiment, receiver 100employs the same edge and data samples used to recover the phase of theincoming data stream to optimize receive equalization. This reuse of theclock and data recovery circuitry for adaptive equalization is powerefficient and adds little complexity.

Receiver 100 includes an equalizer 105 that equalizes an input datasignal VIN from a receive port 107 to produce an equalized data signalVEQ on a single-ended or differential equalizer output port. (As withother designations herein, VIN and VEQ refer both to signals and theircorresponding ports, lines, or nodes; whether a given designation refersto a signal, node, or port will be clear from the context.) Equalizer105 adjusts the magnitude (e.g., voltage and/or current) of at leastsome data symbols in data signal VIN to account for differences in thespectral content of the symbols and symbol patterns. In someembodiments, equalizer 105 selectively adjusts the voltage amplitude ofat least some of the data symbols in data signal VIN, whereas in otherembodiments equalizer 105 selectively adjusts the current used toexpress at least some of the data symbols in data signal VIN. Assumingthe associated channel exhibits a low-pass filter effect, equalizer 105amplifies input data signal VIN using a range of amplification factors,with higher frequency components of VIN being treated to higheramplification factors. In that case, the degree to which equalizer 105amplifies higher frequency signals relative to lower frequency signalscan be adjusted via an equalizer control port 106 coupled to control busEQ[3:0].

Each of a data sampler 110 and an edge sampler 115 samples the equalizeddata signal VEQ from equalizer 105 in time with respective data and edgeclocks CKD and CKE, which clock recovery circuitry 117 derives from alocal bit-rate reference clock signal CKREF. Data sampler 110 issues aseries of data samples on a sampler output port SA_D[C], where “C” isfor “current” sample, whereas edge sampler 115 issues a series of edgesamples on a second sampler output port SA_EDGE.

Equalizer control circuitry 120 is coupled to the control port ofequalizer 105, and to the outputs of data and edge samplers 110 and 115via a retimer 122 that re-times both edge and data samples into the CKDtime domain. The retimed versions of the data and edge samples areconveyed on respective data and edge ports D[C] and EDGE. As detailedbelow, control circuitry 120 determines whether edge transitionsassociated with specific incoming data patterns are early or late withrespect to edge clock CKE. Control circuitry 120 then adjusts equalizer105 to align those edge transitions with the edge clock. To this end,control circuitry 120 includes a bit correlator 130 that issues a matchsignal MATCH in response to a specific data pattern or patterns, a phasedetector 135 that issues relative-timing signals L (late) and E (early)indicative of whether edge transitions of incoming bits are late orearly, respectively, with respect to edge clock CKE, and someequalization logic 140 that issues control settings to equalizer 105 inresponse to the relative-timing signals and the match signal. The earlyand late signals E and L from corresponding relative-timing ports ofphase detector 135 are also conveyed to clock recovery circuitry 117 foruse in aligning edge and data clock signals CKE and CKD with theincoming data signal. Phase detector 135 and clock recovery circuitry117 may be part of conventional clock and data recovery (CDR) circuitryemployed by receiver 100. Other embodiments employ an analog phasedetector to provide the early/late determination. In the depictedembodiment, a clock signal CAP to bit correlator 130, phase detector135, and equalization logic 140 is a gated version of signal CKD, whichallows the CDR and equalizer control circuitry to be run in a burst-modeto save power. The percentage of time the signal CAP is enabled maydepend upon the CDR bandwidth requirements of the system.

Phase detector 135, which may be an Alexander (bang-bang) phasedetector, logically combines the current data sample D[C], the priordata sample D[P], and the edge sample EDGE between the current and priordata samples to determine whether the data edge between the current andprior data samples is early or late with respect to edge clock CKE.Alexander phase detectors are well known to those of skill in the art,so a detailed discussion is omitted. Briefly, samples D[C], D[P], andEDGE are re-timed versions of the incoming data signal. Samples D[C] andD[P] are one bit period (one unit interval) apart, and sample EDGE issampled at half the bit period between samples D[C] and D[P]. If thecurrent and prior samples D[C] and D[P] are the same (e.g., bothrepresent logic one), then no transition has occurred and there is no“edge” to detect. In that case, the outputs E and L of phase detector135 are both zero. If the current and prior samples D[C] and D[P] aredifferent, however, then the edge sample EDGE is compared with thecurrent and prior samples D[C] and D[P]: if sample EDGE equals priorsample D[P], then late signal L is asserted; and if sample EDGE equalscurrent sample D[C], then the early signal E is asserted. In thisdisclosure, a “late” data edge arrives late with respect to the samplingclock, whereas an “early” data edge arrives early with respect to thesampling clock.

Bit correlator 130 employs a series of flip-flops and a pattern mask 150to determine whether the series of symbols expressed by the incomingdata signal matches a specified pattern. Bit correlator 130 requires anexact match, but this may not be required in every embodiment. Forexample, some embodiments allow the user to specify a mask bit patternthat allows the exclusion of bit positions within the pattern from beingincluded in identifying matches.

FIGS. 2 and 3 depict unequalized input data signals for two datapatterns, (10101) and (11101), respectively, to illustrate how differentdata patterns suffer different distortion. In FIG. 2 , the data pattern200 of alternating ones and zeroes is symmetrical above and below zeroamplitude. The zero crossing 205 is centered on a rising edge of edgeclock CKE between data sample instants 210 and 215. In FIG. 3 , the datapattern 300 that begins with a string of ones is asymmetrical about thezero-amplitude axis. The zero crossing 305 is consequently offset withrespect to a rising edge of edge clock CKE between data sample instants310 and 315. The distortion greatly reduces the area of pattern 300below the zero-amplitude line at sample instant 310, and thus increasesthe probability of a sample error. The equalization methods and circuitsdetailed herein recognize that edge and data distortion are patternspecific and focus on reducing the edge distortion of the mosttroublesome data pattern or patterns.

FIG. 4 is a flowchart 400 describing an adaptive equalization methodcarried out by receiver 100 of FIG. 1 in accordance with one embodiment.In this example, pattern mask 150 may be assumed to be configured tocompare incoming data to pattern (11101), the pattern of waveform 300 ofFIG. 3 . Waveform 300 is reproduced in FIG. 5 to illustrate the impactof equalization on pattern 300.

Beginning at step 405, receiver 100 begins receiving data. Theflip-flops within bit correlator 130 provide the incoming data patternsto pattern mask 150. Per decision 410, in the event the incoming datamatches the selected pattern (11101), bit correlator 130 asserts matchsignal MATCH; otherwise, control circuitry 120 simply awaits the nextdata bit.

If ever the incoming data and mask bits do match, equalization logic 140considers the early and late signals L and E from phase detector 135(decisions 415 and 420). If the current and prior data symbols D[C] andD[P] are the same, there is no edge to be late or early, so decisions415 and 420 merely return the flow to step 405 to await the next datasample. If late signal L is asserted, however, equalization logic 140increments a counter (step 430) that filters the “yes” signals fromdecision 415. If the counter of step 430 saturates high (decision 435),equalization logic 140 resets the counter and decrements equalizationcontrol signal EQ[3:0] on the like-named control bus (step 440) toreduce the amplification factor imposed on high-frequency signalcomponents.

If late signal L is not asserted for decision 415, the process flowmoves to decision 420 for consideration of early signal E. If earlysignal E is asserted, equalization logic 140 decrements the counternoted above in connection with step 430 (step 445). If the countersaturates low (decision 450), equalization logic 140 resets the counterand increments equalization control signal EQ[3:0] (step 455) toincrease the amplification factor imposed on high-frequency signalcomponents. Of interest, the flow should not proceed from decision 420directly to block 405 in the example in which a match requires pattern(11101) because that pattern always includes a transition, zero-to-one,between the last two bits.

Turning to the example of FIG. 5 , waveform 300 corresponds to the bitpattern specified by pattern mask 150, and exhibits an edge with a zerocrossing 500 that is early relative to the rising edge clock at instant505. Per FIG. 4 , the combination of an early zero crossing and amatching data pattern causes equalization control signal EQ[3:0] toincrement. The process will continue, with the equalization controlsignal changing each time the incoming data pattern matches the mask bitpattern until the equalized waveform exhibits a zero crossingsynchronized to edge clock CKE. Such a synchronized waveform 505, withzero crossing 510, is depicted in FIG. 5 .

Applicant has discovered that, in certain circumstances, equalizingincoming signals to minimize the timing error associated with theworst-case data pattern or patterns may provide suitable timing forless-sensitive data patterns. Setting the equalizer based upon theworst-case data pattern or patterns therefore causes incoming signalsassociated with various data patterns to exhibit coincident zerocrossings, as is desired. FIG. 6 depicts waveforms 200 and 500, of FIGS.2 and 5 respectively, overlaid to show coincident zero crossings 600 forthe two waveforms. The example of FIG. 6 shows the convergence of justtwo waveforms for ease of illustration. The consideration of additionaldata patterns and associated waveforms may provide better signalconvergence in other embodiments.

In the example of FIG. 4 , the decision to increment or decrementequalization signal EQ[3:0] is based upon that pattern (11101), in whichcase a number of early matches should result in an increase in signalEQ[3:0] and a number of late matches should result in a decrease. Otherpatterns would require the opposite equalization signal adjustment forearly and late matches, however. Early matches for pattern (00001) mightrequire a reduction in equalization signal EQ[3:0], for example, insteadof an increase. Equalization logic 140 can therefore be adapted toprovide pattern-specific feedback to equalizer 105. One such embodimentis detailed below in connection with FIGS. 14 and 15 .

FIG. 7 depicts a double-data-rate (DDR) communication system 700 inaccordance with another embodiment. System 700 includes a transmitter705 that transmits a differential input data signal Vin (Vin_p/Vin_n) toa receive port of a receiver 710 via a differential channel 715. Aconventional transmitter may be employed as transmitter 705, so adetailed treatment is omitted here for brevity. Transmitter 705optionally includes transmit pre-emphasis circuitry (not shown) todynamically adjust data signal Vin to reduce signal distortion caused bythe effects of channel 715. Such transmit pre-emphasis circuitry mayinclude, for example, a multi-tap transmit amplifier adapted to causethe voltage amplitudes of the data symbols of signal Vin to beselectively increased or decreased based of the data values of preand/or post cursor data symbols. Transmitter 705 is typically part of alarger integrated circuit (IC) 720. Although not shown, if a suitablebackchannel is provided between receiver 710 and transmitter 705 thepattern-specific feedback methods and circuitry of receiver 710 can beused to adaptively optimize transmit preemphasis.

Receiver 710 includes an equalizer 725 that equalizes data signal Vin toproduce an equalized data signal VEQ. Equalizer 725 adjusts themagnitude of at least some data symbols in data signal VIN. In oneembodiment, equalizer 725 amplifies signal VIN using a range ofamplification factors, with higher frequency components being treated tohigher amplification factors to, for example, compensate for thelow-pass nature of channel 715. The degree to which equalizer 725amplifies higher frequency signals relative to lower frequency signalscan be adjusted via an equalizer control bus EQ[3:0]. In this example, acontrol port of equalizer 725 receives analog control signals derivedfrom signal EQ[3:0] by a digital-to-analog converter (DAC) 727, thoughequalizer 725 may respond directly to digital signals in otherembodiments.

In support of DDR operation, two data samplers 730 and 732 sample oddand even data symbols using data clocks CK0 and CK180, respectively, toproduce corresponding odd and even data samples DO and Dl. A pair ofedge samplers 735 and 737 likewise sample odd and even edges using edgeclocks CK90 and CK270, respectively, to produce odd and even edgesamples E0 and E1. A deserializer 740 combines the odd and even datasamples into N+1 bit data words DA[N:0], and further combines the oddand even edge samples into N+1 bit edge words ED[N:0]. Data wordsDA[N:0] are conveyed into the integrated circuit as the received datavia a register 742. For both edge and data samples, the bit “N”corresponds to the eldest serial sample and the bit “0” refers to themost recent.

Deserializer 740 is timed, in this embodiment, using a data clock signalDCLK and a word-clock signal WDCLK. Signal DCLK is derived from the sameDDR clock source used to generate clock signals CK0, CK90, CK180, andCK270. The purpose of signal DCLK is to retime all the data and edgesamples into the same time domain and to operate the deserializer. Datais clocked out of deserializer 740 using word-clock signal WDCLK.Register 742 downstream from deserializer 740 is also timed toword-clock signal WDCLK.

Receiver 710 includes equalizer control circuitry 745 that adjustsequalization control signal EQ[3:0] based upon phase information derivedfrom data and edge words DA[N:0] and ED[N:0] for specific data patterns.In accordance with the depicted embodiment, phase information is onlyconsidered when the incoming data word exhibits one or more particulardata patterns PATT[4:0]. As in the foregoing examples, the data patternor patterns may be selected such that correcting for phase errorassociated the selected pattern or patterns produces a suitableequalization setting for all expected data patterns.

Control circuitry 745 includes an Alexander phase detector 750, apattern mask 755, and equalization control logic 760. Phase detector 750identifies data edges by comparing adjacent data samples and, whereedges occur, compares the sampled edge between the adjacent samples todetermine whether the edge is early or late. Phase detector 750 thenprovides, based upon these determinations, an N-bit late wordLATE[N−1:0] and an N-bit early word EARLY[N−1:0]. Pattern mask 755compares each five-bit subset of each data word DA[N:0] with five-bitpattern PATT[4:0] and identifies matches by asserting signal MATCH.Control logic 760 then adjusts equalization control signal EQ[3:0] whena counter (not shown) saturates high or low, thus indicating that matchsignal MATCH is occurring more frequently in conjunction with eitherearly signals or late signals. Phase detector 750 may be part of theclock-recovery circuitry (not shown) used to derive clock signals CK0,CK90, CK180, and CK270. The selected pattern is loaded via a pattern busPATT[4:0] upon assertion of a load signal PLOAD.

FIG. 8 depicts pattern mask 755 and equalization logic 760, both of FIG.7 , in accordance with one embodiment. Pattern mask 755 loads each dataword DA[N:0] into a register 800 upon assertion of a load signal LOAD.Pattern-matching logic 805 compares the first five-bit series inregister 800 (bits D[4:0]) to a selected five-bit pattern PATT[4:0]previously loaded into a pattern register 810. A shift signal SHIFT isperiodically asserted to shift the data in register 800 to drop the mostrecent data sample, thus presenting the next five-bit series topattern-matching logic 805. Pattern-matching logic 805 asserts a matchsignal MATCH when a match is encountered. As in the previous example,the selected pattern is (11101), though another pattern or other sets ofpatterns might also be used. A multi-pattern embodiment is detailedbelow in connection with FIGS. 14 and 15 .

Equalization logic 760 includes early and late registers 815 and 820,into which are loaded N early bits and N late bits, respectively. Aswith data word register 800, the early and late bits are loaded andshifted when the signals LOAD and SHIFT are asserted. The most recentearly and late bits in respective registers 815 and 820 output as earlyand late signals EARLY and LATE, respectively.

Equalization logic 760 additionally includes an XOR gate 825, a NANDgate 830, an AND gate 835, a saturation counter 840, and a counter 845.XOR gate 825 asserts a phase error signal PH ERR when either of theearly and late signals is asserted. AND gate 835 asserts an enablesignal ENAB when pattern mask 755 identifies a match and the early andlate registers indicate a phase error. The enable signal ENAB allowssaturation counter 840 to increment or decrement. NAND gate 830 combinesthe early and late signals EARLY and LATE such that signal UPDN isasserted when early signal EARLY is asserted and late signal LATE isdeasserted, and is deasserted when early signal EARLY is deasserted andlate signal LATE is asserted. Asserting (de-asserting) signal UPDNcauses counter 840 to increment (decrement) when enabled. Counter 840thus counts up or down, depending upon the timing of a zero crossing,when the pattern of received data matches a selected pattern.

FIG. 9 is a waveform diagram 900 illustrating the operation of patternmask 755 and the early and late registers 815 and 820 upon receipt of aten-bit data word (0101011101) extracted from a received waveform 905(i.e., DA[9:0]=(0101011101), where DA9 is the eldest bit). The eldestbit is depicted at the far left of diagram 900, so the binary numberDA[9:0] should be read from right to left. Clock signal CKE combines thesample instants of the two edge clock signals CK90 and CK270 toillustrate edge-sample timing.

The first half of waveform 905 is alternating ones and zeros, and thefirst five zero crossings 910 are close to the corresponding risingedges of clock signal CKE. The sampled zero crossings for suchrelatively symmetrical waveforms can be expected to fluctuate betweenearly and late due to random and periodic jitter sources. The latterhalf of waveform 905 exhibits a series of ones with an intervening zero,which produces a pair of edges for which the zero crossings 915 and 920are relatively far from the corresponding edges of clock signal CKE. Theoutputs of phase detector 750 (FIG. 7 ) are binary, so the early andlate data do not differentiate between edge samples that are slightlyoff and those that are relatively far off. Recall also that both earlyand late signals are zero when no transition is detected (i.e., when twoadjacent data symbols have the same value). As depicted in FIG. 9 , theearly and late words for the hypothetical waveform 905 are, with bitsread from left (eldest) to right (newest): EARLY[8:0]=(101110001); andLATE[8:0]=(010000010).

The following Table 1 illustrates the examination of the data wordDA[9:0]=(0101011101) of FIG. 9 by pattern mask 755 and the early andlate registers 815 and 820 to determine whether to adjust equalizationsignal EQ[3:0]. The ten-bit data word has six unique five-bit series,which are underlined in respective rows zero to five in Table 1. Serieszero, the five most recent bits (11101), matches the selected pattern ofthis example. Pattern mask 755 thus asserts match signal MATCH(MATCH=1). The early bit corresponding to the last transition of (11101)is asserted and the late bit deasserted (the underlined early and latebits of Series zero in Table 1), so signal UPDN is asserted (UPDN=1).AND gate 835 also asserts enable signal ENAB because both match signalMATCH and phase-error signal PH ERR are asserted. Saturation counter 840therefore increments. The five-bit series underlined in each row ofcolumn two in Table 1 corresponds to the bits D[4:0] presented topattern-matching block 805.

TABLE 1 SE- EARLY LATE RIES D[4:0] MATCH [8:0] [8:0] ENAB UPDN 00101011101 1 101110001 010000010 1 1 1 0101011101 0 101110001 0100000100 0 2 0101011101 0 101110001 010000010 0 1 3 0101011101 0 101110001010000010 0 1 4 0101011101 0 101110001 010000010 0 1 5 0101011101 0101110001 010000010 0 1

Each remaining five-bit series is likewise examined, with the shift fromone series to the next taking place upon assertion of shift signalSHIFT. In practice, the data bits may be shifted out of register 800 onebit at a time to present each successive bit series to pattern matchinglogic 805. The first three early and late bits need not be considered,and may therefore be discarded. The early and late bits may be shiftedout of their respective registers 815 and 820 one bit at a time, againupon assertion of signal SHIFT, to present the appropriate early andlate bits EARLY and LATE for consideration in connection with thecorresponding bit pattern. The five series numbered one through five donot result in pattern matches, so the count in counter 840 is unaffecteddespite the value of signal UPDN.

Shifting of the mask pattern is optional. Other embodiments match all ora subset of captured data words, ignoring unconsidered bits. In theforegoing embodiments, the shifting increases the amount of early/lateinformation obtained for a given quantity of sample data and edge data.

FIG. 10 is a flowchart 1002 depicting a method of operation for thecircuitry of FIG. 8 in accordance with one embodiment. Before initiatingadaptive equalization, a pattern is selected to be the basis forequalization adaptation (step 1005). The pattern may be stored locally,e.g. in a volatile or non-volatile memory. Next, at step 1007,saturation counter 840 is set midrange, or binary 100 in this example.

Block 1015 represents the receipt of incoming data and edge samples. Asnoted above in connection with FIG. 8 , the adaptation control circuitryreceives data and edge samples in N+1 bit words. Each five-bit serieswithin each data word is then compared to the selected pattern, whilethe early and late signals associated with the last signal transition ofthe five-bit series are considered to detect a phase error. Per decision1020, if the current five-bit series matches the selected pattern, andthe early and late signals indicate a phase error, then the method flowmoves to decision 1025; otherwise, the process returns to block 1015 forconsideration of the next five-bit series of data. The pattern selectedto adjust equalization will exhibit an edge for comparison to the clocksignal. Otherwise, the selected pattern will not provide an edgedetection upon which to base a phase-error measurement. There shouldtherefore be a phase-error signal whenever the selected pattern isencountered. XOR gate 825 and AND gate 835 are nevertheless included toexclude possible error states in which both early and late signals areone or zero in response to a data edge. Other embodiments omit gates 825and 835, while still other embodiments include phase detectors designedto prevent the above-mentioned error states. Moreover, pattern mask 755may not require an exact match to issue the match signal in someembodiments. Pattern mask 755 might, for example, assert match signalMATCH if at least three of the preceding five bits are a logic one.Gates 825 and 835 would then allow updates to saturation counter 840only if the received pattern included a transition between the final twobits.

If phase error signal PH ERR is asserted, then the sampled edge wasdetermined to be either early or late. In decision 1025, if early thezero crossing was determined to be early (i.e., if signal EARLY isasserted), then saturation counter 840 is incremented (step 1030). Ifthe phase error is not due to an early signal, than late signal LATE wasasserted, in which case saturation counter 840 is decremented (step1035).

Considering next decision 1040, saturation counter 840 may reach themaximum count upon incrementing. If so, counter 840 asserts signalSAT_HI, the upper threshold, in which case equalization control logic845 increments equalization control signal EQ[3:0] and counter 840 isreset to the midrange (steps 1042 and 1007). If counter 840 is notsaturated, then the process returns to block 1015 for consideration ofthe next five-bit series of incoming data. Decision 1045 is similar todecision 1040, except that counter 845 is decremented (step 1050) ifcounter 840 asserts signal SAT_LO, the lower threshold. The upper andlower thresholds of saturation counter 840 may be adjusted in someembodiments.

The phase detector will periodically indicate early and late edges evenwhen the equalization setting is optimal for a given operationalenvironment. Assuming proper data alignment, the resulting early andlate signals will tend to cancel each other over time. That is, counter840 will drift between the upper and lower threshold levels withoutasserting either of signals SAT_HI or SAT_LO. When the edges of theincoming data signal are consistently misaligned with the edge clock,however, the errors will accumulate over time and will consequentlyproduce appropriate counteracting adjustments to equalization controlsignal EQ[3:0]. Saturation counter 840 and equalization control logic845 thus act as a low-pass filter for updates to the equalizationsettings.

FIGS. 11 and 12 are waveform diagrams depicting a pair of data waveforms1105 and 1115 representing respective received data patterns (11101) and(00001). FIG. 11 shows waveforms 1105 and 1115 prior to equalizationadjustment. The zero crossings 1120 and 1125 of waveforms 1105 and 1115are respectively early and late compared with edge clock sample instant1127. Post calibration, depicted in FIG. 12 , the zero crossings 1200 ofboth equalized waveforms 1105 and 1115 are aligned with respect to theedge clock sample instant 1205.

The two patterns of FIGS. 11 and 12 include transitions preceded by astream of like bits. These patterns are therefore apt to provideoutlying edge-sample timing, and may therefore provide appropriatemeasurements for optimizing the equalization setting. Because bothpatterns 1105 and 1115 are potentially troublesome, the associatedpatterns (11101) and (00001) can be used to determine the optimumequalization setting. In one embodiment, for example, equalization isadjusted to find the setting at which the zero crossing of these twodata patterns most nearly converge on the sample instant. In otherembodiments, the method of FIG. 10 may be repeated for severalpotentially troublesome patterns or sets of patterns. Subsequentequalization adaptation can then be based upon the pattern or set ofpatterns that provides the best performance, such as the lowest biterror rate, or the lowest power usage in achieving a minimum level ofspeed performance. The channel transfer function may be used todetermine the bits in the selected pattern or patterns. Sufficientlylong patterns may be used, for example, for echo cancellation.Furthermore, the bits under consideration need not be sequential.

Returning to the example of FIG. 11 , there may be no equalizationsetting at which zero crossings 1120 and 1125 are coincident with theedge clock. In that case, an equalization setting may be selected suchthat zero crossings 1120 and 1125 are as nearly coincident as theequalizer permits, or are as nearly coincident as the equalizer permitswith the limitation that one edge be early and the other late withrespect to the edge clock.

FIG. 13 depicts a DDR receiver 1300 in accordance with anotherembodiment. Receiver 1300 includes an equalizer 1325 that equalizes adifferential data signal Vin_p/Vin_n to produce an equalized signal VEQ.Equalizer 1325 is digitally controlled by an equalization control signalEQ[3:0] that, in this example, allows for selection of sixteen uniqueamplification factors that compensate for the low-pass nature of anincoming channel 1315.

In support of DDR operation, two data samplers 1330 and 1332 sample oddand even data symbols using data clocks CK0 and CK180, respectively, toproduce corresponding odd and even data samples DO and Dl. A pair ofedge samplers 1335 and 1337 likewise sample odd and even edges usingedge clocks CK90 and CK270, respectively, to produce odd and even edgesamples E0 and E1. A pair of deserializers 1340 and 1345 combine the oddand even data samples into N+1 bit data words DA[N:0], and furthercombine the odd and even edge samples into N+1 bit edge words ED[N:0].Data words DA[N:0] are conveyed into the integrated circuit as thereceived data.

Receiver 1300 includes adaptation control circuitry 1350 that adjustsequalization control signal EQ[3:0] based upon phase information derivedfrom data and edge words DA[N:0] and ED[N:0]. Control circuitry 1350includes a state machine 1355, a pattern register 1360, a data-sampleregister 1363, an edge-sample register 1365, and an equalization-settingregister 1370. Adaptation control circuitry 1350 employs these elementsto provide the functionality of control circuitry 745 of FIG. 7 . Statemachine 1355 may be a finite state machine dedicated to provide thefunctionality detailed herein, or may be a general-purpose processorprogrammed or configured to provide the requisite functionality. In oneembodiment, for example, state machine 1355 is a Real Time InterfaceCo-processor (MC) programmed to consider the data and edge samples ofregisters 1363 and 1365 and a selected pattern or patterns in register1360 to set the contents of register 1370, and thereby establish theequalization setting for equalizer 1325. State machine 1355 communicateswith registers 1360, 1363, 1365, and 1370 via a common bus CAD BUS inthis embodiment, though other embodiment may use one or more dedicatedports.

FIG. 14 depicts pattern mask 755 of FIG. 7 and some equalization logic1400 that together adjust equalization signal EQ[3:0] responsive to upto four mask patterns. The circuitry of this embodiment is similar tothat of FIG. 8 , like-numbered elements being the same or similar.Equalization logic 1400 can track MATCH indications for four patternsusing respective counters 1410-1413. Counters 1410-1413 are selected bye.g. one-hot encoding on a select bus CSEL[3:0] gated via AND logic1415.

In a typical example, a set of patterns for a low-pass channel isselected to reduce timing error in the rising and falling edges of a“lone 1” and “lone 0,” such as those associated with the patterns(00001), (11110), (00010), and (11101). The number of bits in theselected patterns may be determined by the magnitude of attenuation athalf of the bit frequency relative to the attenuation at DC: in general,more attenuation requires more pattern bits. For example, five-bitpatterns may be appropriate for an attenuation of about −10 dB at halfof the bit frequency relative to DC. Other patterns or sets of patternsmay be required by systems with fixed training sequences or encodings,or by systems that employ channels with more complicated frequencyresponses. An appropriate set of patterns can be determined for a rangeof complex channels through simulation; alternatively, an in-situexploration of the pattern space can be performed once the hardware isavailable. In the present example, all five-bit patterns could beconsidered, giving up to 32 patterns.

To begin with, a first of the selected patterns is loaded from thepattern bus PATT[4:0] into pattern register 810 upon the assertion of apattern-load signal PLOAD. One of counters 1410-1413 is then selected byasserting one bit of select signal CSEL[3:0]. XOR 825 qualifies theresulting MATCH signals by requiring that only one of the EARLY or LATEsignals must be asserted for a legal match. NAND gate 830 asserts thesignal UPDN to instruct the selected counter to increment or decrementwhen ENAB is asserted. For a given pattern match, if LATE=1 and EARLY=0,the selected counter will decrement when count signal CNT is toggled; ifLATE=0 and EARLY=1, the selected counter will increment when countsignal CNT is toggled.

In this example, we wish to equalize the incoming signal using fourpatterns (00001), (11110), (11101), and (00010). First, pattern (00001)is loaded into pattern register 810, and select signal CSEL[3:0] is setto (0001) to select saturation counter 1410. All five-bit patterns inD[N:0] from register 800 are sequentially compared against the storedmatch pattern in register 810 by toggling SHIFT. Count signal CNT istoggled each time the pattern is shifted, to record whether early signalEARLY or late signal LATE is asserted for the selected pattern.

Next, the pattern (11110) is loaded into pattern register 810 and selectsignal CSEL[3:0] is set to (0010) to select counter 1411. The patternmatching process is then repeated as described above. Likewise, theprocess is repeated for pattern (11101) and counter 1412 and pattern(00010) and counter 1413. Over time, counters 1410-1413 will thus recordthe correlation of early and late indications for the four targetpatterns.

When high-frequency components of a data signal are attenuated more thanthe low-frequency components, counters 1410 and 1411 will tend tosaturate low, and counters 1412 and 1413 will tend to saturate high.Such a state indicates that the equalizer should boost thehigh-frequency gain relative to the low-frequency gain. Conversely, whenthe low-frequency components of a data signal are attenuated more thanthe high-frequency components, counters 1410 and 1411 will tend tosaturate high, and counters 1412 and 1413 will tend to saturate low.Such a state indicates that the equalizer should reduce thehigh-frequency gain relative to the low-frequency gain.

Counters 1410-1413 indicate their respective states to a decoder 1420via respective lines HI_# and LO_#. Adjustment of equalization signalEQ[3:0], and consequently the equalizer, based on the state ofsaturating counter outputs is controlled by a counter decode block 1420.Block 1420 takes as inputs an 8-bit word constructed from the HI and LOoutputs from the four saturating counters 1410-1413 and a control wordEQ_CNTL[7:0]. When the word from counters 1410-1413 equals control wordEQ_CNTL[7:0], decoder 1420 asserts up/down signal UP/DN and enablesignal EQEN, causing an equalization set counter 1425 to increment thebinary value EQ[3:0] on the falling edge of count signal CNT. When theword from counters 1410-1413 equals the inversion of control wordEQ_CNTL[7:0], decoder 1420 deasserts up/down signal UP/DN and assertsenable signal EQEN, causing counter 1425 to decrement EQ[3:0] on thefalling edge of count signal CNT. An AND gate 1430 generates a resetpulse CNT RESET whenever the counter 1425 changes state. This pulseresets the saturating counters to their central values, making themready to measure the effect of the equalization adjustment.

FIG. 15 is a flowchart 1500 depicting the operation of equalizationlogic 1400 of FIG. 14 in accordance with one embodiment. The logic offlowchart 1500 is described here in connection with FIG. 14 .

Beginning with step 1505, registers 1410-1413 and 1425 are each set tothe middle of their ranges and equalization control word EQ_CNTL[7:0] isset to some desired value. Next, at step 1510, the series pointer (notshown) is set to (000) and pattern-select signal CSEL[3:0] is set to(0001). The series pointer selects the first of six five-bit series tobe selected as bits D[4:0] from register 800. Examples of these seriesare detailed above in connection with Table 1. Setting select signalCSEL[3:0] to (0001) enables saturation counter 1410. The series pointermay encode an offset for the selection of the five-bit series fromregister 800, treating register 800 like a memory element rather than ashift register. In some embodiments, register 800 could be a bank ofrandom-access memory (RAM).

In step 1515 register 800 captures N data bits, early register 815captures N−1 early bits, and late register 820 captures N−1 late bits,where N is e.g. ten. Decision 1520 next considers whether the data inregister 800 has been compared with all of the data patterns underconsideration, four in the present example. As no patterns have yet tobe considered, the flow passes to step 1525 in which one of the fourselected patterns is loaded into pattern register 810. Recalling that,from step 1510, the series pointer is selecting the first (most recent)five-bit series in register 800, decision 1530 determines whether thatfive-bit series matches the selected pattern and the registers 815 and820 are producing just one early or late signal. If not, the processmoves to decision 1535 to determine whether all of the five-bit seriesin register 800 have been considered. In this example in which there aresix such series to be considered, decision 1535 increments the seriespointer (step 1545) so long as the series pointer is less than (101). Ifall the series have been considered, however, the process moves to step1540 to enable the next one of saturation counters 1410-1413 inanticipation of comparing the captured data word to another maskpattern.

Returning to decision 1530 and assuming a match with attending phaseerror, the process moves to decision 1550. AND gate 830 drives signalUPDN high to increment the enabled one of saturation counters 1410-1413(step 1555) if signals EARLY and LATE indicate receipt of an early edge;conversely, AND gate 830 drives signal UPDN low to decrement the enabledone of saturation counters 1410-1413 (step 1560) if signals EARLY andLATE indicate receipt of a late edge.

Over time, the state of saturation counters 1410-1413 may match theselected equalization-control word EQ_CNTL[7:0]. Per decision 1565, sucha match causes register 1425 to increment the value of signal EQ[3:0](step 1570). Absent such a match, per decision 1575, the state of thesaturation counters is compared with the inverse of the selectedequalization-control word EQ_CNTL[7:0], which may bedenoted/EQ_CNTL[7:0]. Should such a match occur, register 1425decrements the value of signal EQ[3:0] (step 1580). If neither ofdecisions 1565 and 1575 identifies a match the process moves to decision1535. If a match is noted, however, the resulting increment or decrementis followed by toggling signal CNT RST high then low (step 1585), whichreturns each of saturation counters 1410-1413 to their midrange value.

In the foregoing description and in the accompanying drawings, specificterminology and drawing symbols are set forth to provide a thoroughunderstanding of the present invention. In some instances, theterminology and symbols may imply specific details that are not requiredto practice the invention. For example, the interconnection betweencircuit elements or circuit blocks may be shown or described asmulti-conductor or single conductor signal lines. Each of themulti-conductor signal lines may alternatively be single-conductorsignal lines, and each of the single-conductor signal lines mayalternatively be multi-conductor signal lines. Signals and signalingpaths shown or described as being single-ended may also be differential,and vice-versa. Similarly, signals described or depicted as havingactive-high or active-low logic levels may have opposite logic levels inalternative embodiments. As another example, circuits described ordepicted as including metal oxide semiconductor (MOS) transistors mayalternatively be implemented using bipolar technology or any othertechnology in which a signal-controlled current flow may be achieved.With respect to terminology, a signal is said to be “asserted” when thesignal is driven to a low or high logic state (or charged to a highlogic state or discharged to a low logic state) to indicate a particularcondition. Conversely, a signal is said to be “deasserted” to indicatethat the signal is driven (or charged or discharged) to a state otherthan the asserted state (including a high or low logic state, or thefloating state that may occur when the signal driving circuit istransitioned to a high impedance condition, such as an open drain oropen collector condition). A signal driving circuit is said to “output”a signal to a signal receiving circuit when the signal driving circuitasserts (or de-asserts, if explicitly stated or indicated by context)the signal on a signal line coupled between the signal driving andsignal receiving circuits. A signal line is said to be “activated” whena signal is asserted on the signal line, and “deactivated” when thesignal is de-asserted. In any case, whether a given signal is an activelow or an active high will be evident to those of skill in the art.

An output of a process for designing an integrated circuit, or a portionof an integrated circuit, comprising one or more of the circuitsdescribed herein may be a computer-readable medium such as, for example,a magnetic tape or an optical or magnetic disk. The computer-readablemedium may be encoded with data structures or other informationdescribing circuitry that may be physically instantiated as anintegrated circuit or portion of an integrated circuit. Although variousformats may be used for such encoding, these data structures arecommonly written in Caltech Intermediate Format (CIF), Calma GDS IIStream Format (GDSII), or Electronic Design Interchange Format (EDIF).Those of skill in the art of integrated circuit design can develop suchdata structures from schematic diagrams of the type detailed above andthe corresponding descriptions and encode the data structures oncomputer readable medium. Those of skill in the art of integratedcircuit fabrication can use such encoded data to fabricate integratedcircuits comprising one or more of the circuits described herein.

While the present invention has been described in connection withspecific embodiments, variations of these embodiments will be obvious tothose of ordinary skill in the art. As examples,

-   -   1. Embodiments of the invention may be adapted for use with        multi-pulse-amplitude-modulated (multi-PAM) signals; and    -   2. The feedback methods and circuitry may be adapted for use in        other types of equalizers, such as decision-feedback equalizers        (DFEs), including partial-response DFEs.        Moreover, some components are shown directly connected to one        another while others are shown connected via intermediate        components. In each instance the method of interconnection, or        “coupling,” establishes some desired electrical communication        between two or more circuit nodes, or terminals. Such coupling        may often be accomplished using a number of circuit        configurations, as will be understood by those of skill in the        art. Therefore, the spirit and scope of the appended claims        should not be limited to the foregoing description. Only those        claims specifically reciting “means for” or “step for” should be        construed in the manner required under the sixth paragraph of 35        U.S.C. § 112.

What is claimed is:
 1. A receiver comprising: a receive port to receivean input signal expressing a series of input symbols; an equalizercoupled to the receive port to equalize the input signal, responsive toan equalizer control signal, to thereby create an equalized signalexpressing a series of equalized symbols; a data sampler to produce aseries of data samples from the series of equalized symbols; a phasedetector coupled to the data sampler to issue a phase-error signalresponsive to the data samples; a mask circuit coupled to the datasampler to assert a match signal responsive to a pattern of the datasamples; and equalization logic to update the equalizer control signalresponsive to the phase-error signal and the match signal.
 2. Thereceiver of claim 1, further comprising an edge sampler to produce aseries of edge samples from the series of equalized symbols, the phasedetector to issue the phase-error signal responsive to the data samplesand the edge samples.
 3. The receiver of claim 2, further comprisingclock-recovery circuitry coupled to the data sampler and the edgesampler, the clock-recovery circuitry to time the data sampler and theedge sampler responsive to the data samples and the edge samples.
 4. Thereceiver of claim 2, wherein the phase-error signal comprises at leastone of an early signal and a late signal.
 5. The receiver of claim 1,wherein the mask circuit comprises: a pattern register to store thepattern; a data register to store the series of data samples; andpattern-matching logic coupled to the pattern register and the dataregister to assert the match signal responsive to a match between thepattern and the data samples.
 6. The receiver of claim 1, wherein thepattern of the data samples represents a high-frequency component of theinput signal.
 7. The receiver of claim 6, the mask circuit to assert thematch signal responsive to a second pattern of the data samples.
 8. Thereceiver of claim 7, wherein the second pattern represents alower-frequency component of the input signal.
 9. The receiver of claim8, wherein the equalization logic adjusts the equalizer control signalresponsive to a ratio of errors correlated with the high-frequencycomponent to errors correlated with the lower-frequency component. 10.The receiver of claim 1, further comprising: a second data sampler toproduce a second series of data samples from the series of equalizedsymbols; the mask circuit coupled to the second data sampler to assertthe match signal responsive to a second pattern of the second datasamples.
 11. A method comprising: equalizing an input signal expressinga series of symbols to create an equalized signal, the equalizingresponsive to an equalization setting; sampling the equalized signal toproduce a series of data samples; issuing a phase-error signalresponsive to the data samples; asserting a match signal responsive to apattern of the data samples; and adjusting the equalization settingresponsive to the phase-error signal and the match signal.
 12. Themethod of claim 11, further comprising sampling the equalized signal toproduce a series of edge samples and issuing the phase-error signalresponsive to the data samples and the edge samples.
 13. The method ofclaim 12, further comprising sampling the equalized signal responsive toa clock signal and phase adjusting the clock signal responsive to thedata samples and the edge samples.
 14. The method of claim 12, whereinthe phase-error signal comprises at least one of an early signal and alate signal.
 15. The method of claim 11, further comprising: storing apattern; and asserting the match signal responsive to a match betweenthe pattern and the data samples.
 16. The method of claim 11, whereinthe pattern of the data samples represents a high-frequency component ofthe input signal.
 17. The method of claim 16, further comprisingasserting the match signal responsive to a second pattern of the datasamples.
 18. The method of claim 17, wherein the second patternrepresents a lower-frequency component of the input signal.
 19. Themethod of claim 18, further comprising adjusting the equalizationsetting responsive to a ratio of errors correlated with thehigh-frequency component to errors correlated with the lower-frequencycomponent.